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ISL6228
Data Sheet May 7, 2008 FN9095.2
High-Performance Dual-Output Buck Controller for Notebook Applications
The ISL6228 IC is a dual channel synchronous-buck PWM controller featuring Intersil's Robust Ripple Regulator (R3) technology that delivers truly superior dynamic response to input voltage and output load transients. Integrated MOSFET drivers and bootstrap diodes result in fewer components and smaller implementation area. Intersil's R3 technology combines the best features of fixedfrequency and hysteretic PWMs while eliminating many of their shortcomings. R3 technology employs an innovative modulator that synthesizes an AC ripple voltage signal VR, analogous to the output inductor ripple current. The AC signal VR enters a window comparator where the lower threshold is the error amplifier output VCOMP, and the upper threshold is a programmable voltage reference VW, resulting in generation of the PWM signal. The voltage reference VW sets the steady-state PWM frequency. Both edges of the PWM can be modulated in response to input voltage transients and output load transients, much faster than conventional fixedfrequency PWM controllers. Unlike a conventional hysteretic converter, each channel of the ISL6228 has an error amplifier that provides 1% voltage regulation at the FB pin. The ISL6228 has a 1.5ms digital soft-start and can be started into a pre-biased output voltage. A resistor divider is used to program the output voltage setpoint. The ISL6228 operates in continuous-conduction-mode (CCM) in heavy load, and in diode-emulation-mode (DEM) in light load to improve light-load efficiency. In CCM, the controller always operates as a synchronous rectifier. In DEM, the low-side MOSFET is permitted to stay off, blocking negative current flow into the low-side MOSFET from the output inductor.
Features
* High performance R3 technology * Fast transient response * 1% regulation accuracy: -40C to +100C * Individual power stage input rail for each channel * Wide input voltage range: +3.3V to +25V * Output voltage range: +0.6V to +5V * Diode emulation mode for increased light load efficiency * Programmable PWM frequency: 200kHz to 600kHz * Pre-biased output start-up capability * Integrated MOSFET drivers and bootstrap diode * Internal digital soft-start * Power good monitor * Fault protection - Undervoltage protection - Soft crowbar overvoltage protection - Inductor DCR overcurrent protection - Over-temperature protection - Fault identification by PGOOD pull-down resistance * Pb-free (RoHS compliant)
Applications
* General purpose switching buck regulators * PCI express graphical processing unit * Auxiliary power rail * VRM * Network adaptor
Pinout
ISL6228 (28 LD 4x4 TQFN)
PGOOD2 OCSET2 PHASE2 UGATE2 VO2 EN2 FB2
Ordering Information
PART NUMBER PART (Note) MARKING ISL6228HRTZ
21 BOOT2 20 PVCC2 19 LGATE2
TEMP (C)
PACKAGE (Pb-Free)
PKG. DWG. #
28 FSET2 1 VIN2 2 VCC2 3 VCC1 4 VIN1 5 FSET1 6 PGOOD1 7 8 FB1
27
26
25 24
23
22
6228HRTZ -10 to +100 28 Ld 4x4 TQFN L28.4x4A
ISL6228HRTZ-T* 6228HRTZ -10 to +100 28 Ld 4x4 TQFN L28.4x4A Tape and Reel ISL6228IRTZ 6228IRTZ -40 to +100 28 Ld 4x4 TQFN L28.4x4A
GND
18 PGND2 17 PGND1 16 LGATE1 15 PVCC1
ISL6228IRTZ-T* 6228IRTZ -40 to +100 28 Ld 4x4 TQFN L28.4x4A Tape and Reel *Please refer to TB347 for details on reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pbfree material sets; molding compounds/die attach materials and 100% matte tin plate PLUS ANNEAL - e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
9 VO1
10 OCSET1
11 12 EN1 PHASE1
13 UGATE1
14 BOOT1
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2007, 2008. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
Block Diagram
VIN2 FB2 VCC2 VCC1 FB1 VIN1
VO2 FSET2 - PWM FREQUENCY CONTROL + VW - PWM + gm CR - + gm - CR + 10A 10A + gm - PWM FREQUENCY CONTROL + VW - PWM -
VO1 FSET1
R Q S
+ EN2
VCOMP
+ -
+ -
VCOMP
100pF EA POR + + - VREF VREF - EA
100pF
BOOT2
POR DIGITAL SOFTSTART PWM CONTROL DRIVER UGATE1
UGATE2
DRIVER
PHASE2 SHOOT THROUGH PROTECTION
+ UVP -
+ UVP -
+ OCP LGATE2 DRIVER -
10A
10A
+150 OT
60
90
30
PACKAGE BOTTOM
FN9095.2 May 7, 2008
PVCC2
PGND2
PGOOD2
OCSET2
GND
OCSET1
FIGURE 1. SCHEMATIC BLOCK DIAGRAM
-
-
-
DIGITAL SOFTSTART PWM CONTROL
+ OVP
+ OVP
SHOOT THROUGH PROTECTION
+ OCP DRIVER LGATE1
30
90
60
+150 OT
PGOOD1
PGND1
-
VR
-
+
gm
+ R Q S
-
2
VR
+ EN1
BOOT1
ISL6228
PHASE1
PVCC1
ISL6228 Typical Application
5V
RPGOOD2
RPVCC2
RVCC2 VCC2 PVCC2 VCC1 PVCC1
RVCC1
RPVCC1
RPGOOD1
CPVCC2 PGOOD2
CVCC2
CVCC1
CPVCC1 PGOOD1
PGOOD2 VIN2 VIN2 3.3V TO 25V CIN2 QHIGH_SIDE2 UGATE2 BOOT2
PGOOD1 VIN1 VIN1 3.3V TO 25V QHIGH_SIDE1 CIN1
UGATE1 BOOT1
VO2 0.6V TO 5V CO2
LO2
CBOOT2 PHASE2 PHASE1
CBOOT1
LO1
VO1 0.6V TO 5V
CSEN2 ROCSET2
QLOW_SIDE2
ISL6228
QLOW_SIDE1
CSEN1 ROCSET1
CO1
LGATE2 PGND2 OCSET2 RFB2 CFB2 VO2 RO2 FB2 FSET2 RFSET2 CFSET2
LGATE1 PGND1
OCSET1 VO1 RO1 FB1 FSET1 CFSET1 RFSET1 RFB1 CFB1
RTOP2
RTOP1
RBOTTOM2
RBOTTOM1
EN2 GND
EN1
3
FN9095.2 May 7, 2008
ISL6228
Absolute Voltage Ratings
VIN1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V VCC, PGOOD1,2 to GND . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V PVCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V EN1,2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to GND, VCC +3.3V VO1,2, FB1,2, FSET1,2 . . . . . . . . . . . . . . -0.3V to GND, VCC +0.3V PHASE1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . (DC) -0.3V to +28V (<100ns Pulse Width, 10J) . . . . . . . . . . . . . . . . . . . . . . . . . -5.0V BOOT1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V BOOT1,2 to PHASE1,2 . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V UGATE1,2 . . . . . . . . . . . . (DC) -0.3V to PHASE1,2, BOOT1,2 +0.3V (<200ns Pulse Width, 20J) . . . . . . . . . . . . . . . . . . . . . . . . -4.0V LGATE1,2 . . . . . . . . . . . . . . . . . . . (DC) -0.3V to GND, PVCC +0.3V (<100ns Pulse Width, 4J) . . . . . . . . . . . . . . . . . . . . . . . . . . -2.0V
Thermal Information
Thermal Resistance (Typical, Notes 1, 2) JA (C/W) JC (C/W) TQFN Package . . . . . . . . . . . . . . . . . . 40 3 Junction Temperature Range. . . . . . . . . . . . . . . . . .-55C to +150C Operating Temperature Range . . . . . . . . . . . . . . . .-40C to +100C Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65C to +150C Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Ambient Temperature Range. . . . . . . . . . . . . . . . . .-10C to +100C Supply Voltage (VIN to GND) . . . . . . . . . . . . . . . . . . . . 3.3V to 25V VCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5V 5% PVCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5V 5%
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.
NOTES: 1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features. See Tech Brief TB379. 2. For JC, the "case temp" location is the center of the exposed metal pad on the package underside. 3. Limits established by characterization and are not production tested.
Electrical Specifications
These specifications apply for TA = -40C to +100C; All typical specifications TA = +25C, VCC = 5V, PVCC = 5V; Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. SYMBOL TEST CONDITIONS MIN TYP MAX UNIT
PARAMETER VIN VIN Input Bias Current VIN Shutdown Current VCC and PVCC VCC Input Bias Current in Single-Channel VCC Input Bias Current in Dual Channel VCC Shutdown Current PVCC Shutdown Current VCC POR THRESHOLD Rising VCC POR Threshold Voltage
IVIN
EN = 5V, VIN = 15V
16 0.1
1.0
A A
IVIN_SHDN EN = GND, VIN = 25V IVCC_S IVCC_D EN1 = 5V, FB1 = 0.65V, VIN1 = 3.3V to 25V, EN2 = GND, FB2 = GND, VIN2 = GND EN1 = 5V, FB1 = 0.65V, VIN1 = 3.3V to 25V, EN2 = 5V, FB2 = 0.65V, VIN2 = 3.3V to 25V
-
1 2 0.1 0.1
1.0 1.0
mA mA A A
IVCC_SHDN EN1 = GND, EN2 = GND, VCC = 5V IPVCC_SHDN EN1 = GND, EN2 = GND, PVCC = 5V VVCC_THR TA = -10C to +100C V
4.33 4.35 4.08 TA = -10C to +100C 4.10 Close loop -1
4.45 4.45 4.20 4.20
4.55 4.55 4.30 4.30 +1
V V V V
Falling VCC POR Threshold Voltage
VCC_THF
REGULATION Reference Voltage Regulation Accuracy PWM Frequency Range Frequency-Set Accuracy VO Range VO Input Leakage VVO IVO EN = 5V, VO = 0.60V EN = 5V, VO = 5V EN = 0V, VO = 5V fSW fSW = 300kHz 200 -12 0.60 1 7.0 0.1 600 +12 5 kHz % V A A A VREF 0.6 V %
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ISL6228
Electrical Specifications
These specifications apply for TA = -40C to +100C; All typical specifications TA = +25C, VCC = 5V, PVCC = 5V; Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued) SYMBOL TEST CONDITIONS MIN -33 -35 TYP MAX 15 15 UNIT
PARAMETER ERROR AMPLIFIER FB Input Bias Current
IFB
FB = 0.60V FB = 0.60V TA = -10C to +100C
-
nA nA A mA ms A A A A ns ns
POWER GOOD PGOOD Pull-Down Impedance RPG_SS RPG_SS RPG_UV RPG_UV RPG_OV RPG_OV RPG_OC RPG_OC PGOOD Leakage Current PGOOD Maximum Sink Current (Note 3) PGOOD Soft-Start Delay GATE DRIVER UGATE Pull-Up Resistance (Note 3) UGATE Source Current (Note 3) UGATE Sink Resistance (Note 3) UGATE Sink Current (Note 3) LGATE Pull-Up Resistance (Note 3) LGATE Source Current (Note 3) LGATE Sink Resistance (Note 3) LGATE Sink Current (Note 3) UGATE to LGATE Deadtime LGATE to UGATE Deadtime BOOTSTRAP DIODE Forward Voltage Reverse Leakage CONTROL INPUTS EN High Threshold EN Low Threshold EN Leakage VENTHR VENTHF IENL IENH PROTECTION OCSET-VO Threshold OCSET 10A Current Source VOCSETTHR IOCSET EN = 5V EN = 5V TA = -10C to +100C EN = 0V OCSET 10A Current Source Impedance ROCSETIMP EN = 5V, OCSET = 1.2V UVP Threshold OVP Rising Threshold OVP Falling Threshold VUV VOVR VOVF -1.75 8.8 9 81 113 100 0 10 10 0 600 86 116 102 1.75 10.5 10.5 87 120 106 mV A A A k % % % EN = 0V EN = 5.0V EN = 5.0V TA = -10C to +100C 2.0 1.4 1.5 0.1 2 2 1.0 1.0 2.5 2.5 V V A A A VF IR PVCC = 5V, IF = 2mA VR = 25V 0.58 0.2 V A RUGPU IUGSRC RUGPD IUGSNK RLGPU ILGSRC RLGPD ILGSNK tUGFLGR tLGFUGR 200mA Source Current UGATE - PHASE = 2.5V 250mA Sink Current UGATE - PHASE = 2.5V 250mA Source Current LGATE - PGND = 2.5V 250mA Sink Current LGATE - PGND = 2.5V UGATE falling to LGATE rising, no load LGATE falling to UGATE rising, no load 1.0 2.0 1.0 2.0 1.0 2.0 0.5 4.0 21 21 1.5 1.5 1.5 0.9 tSS EN High to PGOOD High IPGOOD PGOOD = 5mA Sink PGOOD = 5mA Sink TA = -10C to +100C PGOOD = 5mA Sink PGOOD = 5mA Sink TA = -10C to +100C PGOOD = 5mA Sink PGOOD = 5mA Sink TA = -10C to +100C PGOOD = 5mA Sink PGOOD = 5mA Sink TA = -10C to +100C PGOOD = 5V 70 75 70 75 45 50 22 25 2.20 95 95 95 95 63 63 32 32 0.1 5.0 2.75 125 125 125 125 85 85 45 45 1.0 3.50
5
FN9095.2 May 7, 2008
ISL6228
Electrical Specifications
These specifications apply for TA = -40C to +100C; All typical specifications TA = +25C, VCC = 5V, PVCC = 5V; Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued) SYMBOL TOTR TOTHYS TEST CONDITIONS MIN TYP 150 25 MAX UNIT C C
PARAMETER OTP Rising Threshold (Note 3) OTP Hysteresis (Note 3)
Functional Pin Descriptions
GND (Bottom Pad)
Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin.
connected across the VO1, FB1, and GND pins. Select the resistor values such that FB1 to GND is 600mV when the converter output voltage is at the programmed regulation value.
VO1 (Pin 9)
The VO1 pin measures the Channel 1 converter output voltage and is used as an input to the Channel 1 R3 PWM modulator. It also serves as part of Channel 1 inductor current sensing and the OCP overcurrent fault protection circuit.
FSET2 (Pin 1)
The FSET2 pin programs the PWM switching frequency of Channel 2. Program the desired PWM frequency with a resistor and a capacitor connected across the FSET2 and GND pins.
VIN2 (Pin 2)
The VIN2 pin measures the input voltage of the Channel 2 converter. It is a required input to the Channel 2 R3 PWM modulator. Connect the VIN2 pin to the drain of the Channel 2 high-side MOSFET.
OCSET1 (Pin 10)
The OCSET1 pin measures the Channel 1 inductor current and programs the threshold of the OCP overcurrent fault protection.
EN1 (Pin 11)
The EN1 pin is the on/off switch of Channel 1. The soft-start sequence begins when the EN1 pin is pulled above the rising threshold voltage VENTHR and VCC1 is above the power-on reset (POR) rising threshold voltage VVCC_THR . When the EN1 pin is pulled below the falling threshold voltage VENTHF PWM1 immediately stops.
VCC2 (Pin 3)
The VCC2 pin is the input bias voltage for Channel 2. Connect +5V to the VCC2 pin. Decouple with at least 1F of a MLCC capacitor from the VCC2 pin to the GND pin.
VCC1 (Pin 4)
The VCC1 pin is the input bias voltage for Channel 1. Connect +5V to the VCC1 pin. Decouple with at least 1F of a MLCC capacitor from the VCC1 pin to the GND pin.
PHASE1 (Pin 12)
The PHASE1 pin is the current return path for the Channel 1 high-side MOSFET gate driver. Connect the PHASE1 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of the Channel 1 converter.
VIN1 (Pin 5)
The VIN1 pin measures the input voltage of the Channel 1 converter. It is a required input to the Channel 1 R3 PWM modulator. Connect the VIN1 pin to the drain of the Channel 1 high-side MOSFET.
UGATE1 (Pin 13)
The UGATE1 pin is the output of the Channel 1 high-side MOSFET gate driver. Connect the UGATE1 pin to the gate of the Channel 1 converter high-side MOSFET.
FSET1 (Pin 6)
The FSET1 pin programs the PWM switching frequency of Channel 1. Program the desired PWM frequency with a resistor and a capacitor connected across the FSET1 and GND pins.
BOOT1 (Pin 14)
The BOOT1 pin stores the input voltage for the Channel 1 high-side MOSFET gate driver. Connect an MLCC capacitor across the BOOT1 and PHASE1 pins. The boot capacitor is charged through an internal boot diode connected from the PVCC1 pin to the BOOT1 pin, each time the PHASE1 pin drops below PVCC1 minus the voltage dropped across the internal boot diode.
PGOOD1 (Pin 7)
The PGOOD1 pin is an open-drain output that indicates when the Channel 1 converter is able to supply regulated voltage. Connect the PGOOD1 pin to +5V through a pull-up resistor.
FB1 (Pin 8)
The FB1 pin is the inverting input of the control-loop error amplifier for Channel 1. The Channel 1 converter output voltage regulates to 600mV from the FB1 pin to the GND pin. Program the desired output voltage with a resistor network 6
PVCC1 (Pin 15)
The PVCC1 pin is the input voltage bias for the Channel 1 low-side MOSFET gate drivers. Connect +5V to the PVCC1
FN9095.2 May 7, 2008
ISL6228
pin. Decouple with at least 1F of an MLCC capacitor across the PVCC1 and PGND1 pin.
EN2 (Pin 24)
The EN2 pin is the on/off switch of Channel 2. The soft-start sequence begins when the EN2 pin is pulled above the rising threshold voltage VENTHR and VCC2 is above the power-on reset (POR) rising threshold voltage VVCC_THR . When the EN2 pin is pulled below the falling threshold voltage VENTHF, PWM2 immediately stops.
LGATE1 (Pin 16)
The LGATE1 pin is the output of the Channel 1 converter low-side MOSFET gate driver. Connect the LGATE1 pin to the gate of the Channel 1 converter low-side MOSFET.
PGND1 (Pin 17)
The PGND1 pin is the current return path for the Channel 1 converter low-side MOSFET gate driver. Connect the PGND1 pin to the source of the Channel 1 converter lowside MOSFET through a low impedance path, preferably in parallel with the trace connecting the LGATE1 pin to the gate of the Channel 1 converter low-side MOSFET.
OCSET2 (Pin 25)
The OCSET2 pin measures the Channel 2 inductor current and programs the threshold of the OCP overcurrent fault protection.
VO2 (Pin 26)
The VO2 pin measures the Channel 2 converter output voltage and is used as an input to the Channel 2 R3 PWM modulator. It also serves as part of Channel 2 inductor current sensing and the OCP overcurrent fault protection circuit.
PGND2 (Pin 18)
The PGND2 pin is the current return path for the Channel 2 converter low-side MOSFET gate driver. Connect the PGND2 pin to the source of the Channel 2 converter lowside MOSFET through a low impedance path, preferably in parallel with the trace connecting the LGATE2 pin to the gate of the Channel 2 converter low-side MOSFET.
FB2 (Pin 27)
The FB2 pin is the inverting input of the control-loop error amplifier for Channel 2. The Channel 2 converter output voltage regulates to 600mV from the FB2 pin to the GND pin. Program the desired output voltage with a resistor network connected across the VO2, FB2, and GND pins. Select the resistor values such that FB2 to GND is 600mV when the converter output voltage is at the programmed regulation value.
LGATE2 (Pin 19)
The LGATE2 pin is the output of the Channel 2 converter low-side MOSFET gate driver. Connect to the gate of the Channel 2 converter low-side MOSFET.
PVCC2 (Pin 20)
The PVCC2 pin is the input voltage bias for the Channel 2 low-side MOSFET gate drivers. Connect +5V to the PVCC2 pin. Decouple with at least 1F of an MLCC capacitor across the PVCC2 and PGND2 pin.
PGOOD2 (Pin 28)
The PGOOD2 pin is an open-drain output that indicates when the Channel 2 converter is able to supply regulated voltage. Connect the PGOOD2 pin to +5V through a pull-up resistor.
BOOT2 (Pin 21)
The BOOT2 pin stores the input voltage for the Channel 2 high-side MOSFET gate driver. Connect an MLCC capacitor across the BOOT2 and PHASE2 pins. The boot capacitor is charged through an internal boot diode connected from the PVCC2 pin to the BOOT2 pin, each time the PHASE2 pin drops below PVCC2 minus the voltage dropped across the internal boot diode.
Theory of Operation
Two Separate Channels
The ISL6228 is a dual channel controller. Pins 4~17 are dedicated to Channel 1, and pins 1~3 and pins 18~28 are dedicated to Channel 2. The two channels are identical and almost entirely independent, with the exception of sharing the GND pin. Unless otherwise stated, only an individual channel is discussed, and the conclusion applies to both channels.
UGATE2 (Pin 22)
The UGATE2 pin is the output of the Channel 2 high-side MOSFET gate driver. Connect to the gate of the Channel 2 converter high-side MOSFET.
Modulator
The ISL6228 modulator features Intersil's R3 technology, a hybrid of fixed frequency PWM control and variable frequency hysteretic control. Intersil's R3 technology can simultaneously affect the PWM switching frequency and PWM duty cycle in response to input voltage and output load transients. The R3 modulator synthesizes an AC signal VR, which is an analog representation of the output inductor ripple current. The duty-cycle of VR is the result of charge and discharge current through a ripple capacitor CR. The
PHASE2 (Pin 23)
The PHASE2 pin is the current return path for the Channel 2 high-side MOSFET gate driver. Connect the PHASE2 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of the Channel 2 converter.
7
FN9095.2 May 7, 2008
ISL6228
current through CR is provided by a transconductance amplifier gm that measures the VIN and VO pin voltages. The positive slope of VR can be written as Equation 1:
V RPOS = ( g m ) ( V IN - V OUT ) C R (EQ. 1)
Soft-Start Delay tSS begins and the output voltage begins to rise. The FB pin ramps to 0.6V in approximately 1.5ms and the PGOOD pin goes to high impedance approximately 1.25ms after the FB pin voltage reaches 0.6V.
1.5ms
The negative slope of VR can be written as Equation 2:
V RNEG = g m V OUT C R (EQ. 2)
Vo
VCC and PVCC
Where gm is the gain of the transconductance amplifier. A window voltage VW is referenced with respect to the error amplifier output voltage VCOMP, creating an envelope into which the ripple voltage VR is compared. The amplitude of VW is set by a resistor connected across the FSET and GND pins. The VR, VCOMP, and VW signals feed into a window comparator in which VCOMP is the lower threshold voltage and VW is the higher threshold voltage. Figure 2 shows PWM pulses being generated as VR traverses the VW and VCOMP thresholds. The PWM switching frequency is proportional to the slew rates of the positive and negative slopes of VR; it is inversely proportional to the voltage between VW and VCOMP.
EN FB
PGOOD 1.25ms
FIGURE 3. SOFT-START SEQUENCE
Ripple Capacitor Voltage CR
Window Voltage VW
The PGOOD pin indicates when the converter is capable of supplying regulated voltage. The PGOOD pin is an undefined impedance if VCC has not reached the rising POR threshold VCCR, or if VCC is below the falling POR threshold VCCF. The ISL6228 features a unique fault-identification capability that can drastically reduce trouble-shooting time and effort. The pull-down resistance of the PGOOD pin corresponds to the fault status of the controller. The PGOOD pull-down resistance is 95 during soft-start or if an UVP occurs, 30 for an OCP, or 60 for OVP.
TABLE 1. PGOOD PULL-DOWN RESISTANCE CONDITION PGOOD RESISTANCE Undefined 90 60 30
Error Amplifier Voltage VCOMP
VCC Below POR Soft-start or Undervoltage Overvoltage
PWM
Overcurrent
MOSFET Gate-Drive Outputs LGATE and UGATE
FIGURE 2. MODULATOR WAVEFORMS DURING LOAD TRANSIENT
Power-On Reset
The ISL6228 is disabled until the voltage at the VCC pin has increased above the rising power-on reset (POR) VCCR threshold voltage. The controller will be disabled when the voltage at the VCC pin decreases below the falling POR VCCF threshold voltage.
EN, Soft-Start and PGOOD
The ISL6228 uses a digital soft-start circuit to ramp the output voltage of the converter to the programmed regulation setpoint at a predictable slew rate. The slew rate of the soft-start sequence has been selected to limit the in-rush current through the output capacitors as they charge to the desired regulation voltage. When the EN pin is pulled above the rising EN threshold voltage VENTHR, the PGOOD 8
The ISL6228 has internal gate-drivers for the high-side and low-side N-Channel MOSFETs. The low-side gate-drivers are optimized for low duty-cycle applications where the lowside MOSFET conduction losses are dominant, requiring a low r DS(ON) MOSFET. The LGATE pull-down resistance is small in order to clamp the gate of the MOSFET below the VGS(th) at turnoff. The current transient through the gate at turn-off can be considerable because the gate charge of a low r DS(ON) MOSFET can be large. Adaptive shoot-through protection prevents a gate-driver output from turning on until the opposite gate-driver output has fallen below approximately 1V. The dead-time shown in Figure 4 is extended by the additional period that the falling gate voltage stays above the 1V threshold. The typical dead-time is 21ns. The high-side gate-driver output voltage is measured across the UGATE and PHASE pins while the low-side gate-driver output voltage is measured across the LGATE and PGND
FN9095.2 May 7, 2008
ISL6228
pins. The power for the LGATE gate-driver is sourced directly from the PVCC pin. The power for the UGATE gatedriver is sourced from a "boot" capacitor connected across the BOOT and PHASE pins. The boot capacitor is charged from a 5V bias supply through a "boot diode" each time the low-side MOSFET turns on, pulling the PHASE pin low. The ISL6228 has an integrated boot diode connected from the PVCC pin to the BOOT pin. detected positive voltage and LGATE was allowed to go high for eight consecutive PWM switching cycles. The ISL6228 will turn off the low-side MOSFET once the phase voltage turns positive, indicating negative inductor current. The ISL6228 will return to CCM on the following cycle after the PHASE pin detects negative voltage, indicating that the body diode of the low-side MOSFET is conducting positive inductor current. Efficiency can be further improved with a reduction of unnecessary switching losses by reducing the PWM frequency. It is characteristic of the R3 architecture for the PWM frequency to decrease while in diode emulation. The extent of the frequency reduction is proportional to the reduction of load current. Upon entering DEM, the PWM frequency makes an initial step-reduction because of a 33% step-increase of the window voltage V W.
tLGFUGR
tUGFLGR
50% UGATE LGATE 50%
Overcurrent Protection
The overcurrent protection (OCP) setpoint is programmed with resistor ROCSET that is connected across the OCSET and PHASE pins.
L DCR PHASE + ISL6228 10A OCSET ROCSET + VROCSET RO VO VDCR CSEN _ IL _ CO VO
FIGURE 4. LGATE AND UGATE DEAD-TIME
Diode Emulation
The ISL6228 implements forced continuous-conductionmode (CCM) at heavy load and diode-emulation-mode (DEM) at light load, to optimize efficiency in the entire load range. The transition is automatically achieved by detecting the output load current. Positive-going inductor current flows from either the source of the high-side MOSFET, or the drain of the low-side MOSFET. Negative-going inductor current flows into the drain of the low-side MOSFET. When the low-side MOSFET conducts positive inductor current, the phase voltage will be negative with respect to the GND and PGND pins. Conversely, when the low-side MOSFET conducts negative inductor current, the phase voltage will be positive with respect to the GND and PGND pins. The ISL6228 monitors the phase voltage, when the low-side MOSFET is conducting inductor current, to determine the direction of the inductor current. When the output load current is greater than or equal to 1/2 the inductor ripple current, the inductor current is always positive, and the converter is always in CCM. The ISL6228 minimizes the conduction loss in this condition by forcing the low-side MOSFET to operate as a synchronous rectifier. When the output load current is less than 1/2 the inductor ripple current, negative inductor current occurs. Sinking negative inductor through the low-side MOSFET lowers efficiency through unnecessary conduction losses. The ISL6228 automatically enters DEM after the PHASE pin has
FIGURE 5. OVERCURRENT-SET CIRCUIT
Figure 5 shows the overcurrent-set circuit. The inductor consists of inductance L and the DC resistance DCR. The inductor DC current IL creates a voltage drop across DCR, given by Equation 3:
V DCR = I L * DCR (EQ. 3)
The ISL6228 sinks 10A current into the OCSET pin, creating a DC voltage drop across the resistor ROCSET, given by Equation 4:
V ROCSET = 10A * R OCSET (EQ. 4)
Resistor RO is connected between the VO pin and the actual output voltage of the converter. During normal operation, the VO pin is a high impedance path, therefore there is no voltage drop across RO. The DC voltage difference between the OCSET pin and the VO pin can be established using Equation 5:
V OCSET - V VO = V DCR - V ROCSET = I L * DCR - 10A * R OCSET (EQ. 5)
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The ISL6228 monitors the OCSET pin and the VO pin voltages. Once the OCSET pin voltage is higher than the VO pin voltage for more than 10s, the ISL6228 declares an OCP fault. The value of ROCSET is then written as Equation 6:
I OC * DCR R OCSET = -------------------------10A (EQ. 6)
Where: - ROCSET () is the resistor used to program the overcurrent setpoint - IOC is the output current threshold that will activate the OCP circuit - DCR is the inductor DC resistance For example, if IOC is 20A and DCR is 4.5m, the choice of ROCSET is ROCSET = 20A x 4.5m/10A = 9k. Resistor ROCSET and capacitor CSEN form an R-C network to sense the inductor current. To sense the inductor current correctly not only in DC operation, but also during dynamic operation, the R-C network time constant ROCSETCSEN needs to match the inductor time constant L/DCR. The value of CSEN is then written as Equation 7:
L C SEN = ----------------------------------------R OCSET * DCR (EQ. 7)
the output voltage transversing the VOVR and VOVF thresholds. The LGATE gate-driver will turn on the low-side MOSFET to discharge the output voltage, protecting the load. The LGATE gate-driver will turn off the low-side MOSFET once the FB pin voltage is lower than the falling overvoltage threshold VOVF for more than 2s. The falling overvoltage threshold VOVF is typically 106%. That means if the FB pin voltage falls below 106% x 0.6V = 0.636V, for more than 2s, the LGATE gate-driver will turn off the lowside MOSFET. If the output voltage rises again, the LGATE driver will again turn on the low-side MOSFET when the FB pin voltage is above the rising overvoltage threshold VOVR for more than 2s. By doing so, the ISL6228 protects the load when there is a consistent overvoltage condition.
Undervoltage Protection
The UVP fault detection circuit triggers after the FB pin voltage is below the undervoltage threshold VUV for more than 2s. The FB pin voltage is 0.6V in normal operation. The undervoltage threshold VUV is typically 86%. That means if the FB pin voltage is below 86% x 0.6V = 0.516V, for more than 2s, an UVP fault is declared, and the PGOOD pin will pull down to 95 and latch-off the converter. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage V VCC_THF.
For example, if L is 1.5H, DCR is 4.5m, and ROCSET is 9k, the choice of CSEN = 1.5H/(9k x 4.5m) = 0.037F. Upon converter startup, capacitor CSEN initial voltage is 0V. To prevent false OCP, a 10A current source flows out of the VO pin during start up, generating a voltage drop on resistor RO, which has the same resistance as ROCSET. When PGOOD pin goes high, the VO pin current source will terminate. When an OCP fault is declared, the PGOOD pin will pull down to 30 and latch off the converter. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF.
Programming the Output Voltage
When the converter is in regulation there will be 0.6V from the FB pin to the GND pin. Connect a two-resistor voltage divider across the VO pin and the GND pin with the output node connected to the FB pin. Scale the voltage-divider network such that the FB pin is 0.6V with respect to the GND pin when the converter is regulating at the desired output voltage. The output voltage can be programmed from 0.6V to 5V. Programming the output voltage is written as Equation 8:
R BOTTOM V REF = V O * -------------------------------------------------R TOP + R BOTTOM (EQ. 8)
Overvoltage Protection
The OVP fault detection circuit triggers after the FB pin voltage is above the rising overvoltage threshold VOVR for more than 2s. The FB pin voltage is 0.6V in normal operation. The rising overvoltage threshold VOVR is typically 116%. That means if the FB pin voltage is above 116% x 0.6V = 0.696V, for more than 2s, an OVP fault is declared. When an OVP fault is declared, the PGOOD pin will pull down to 60 and latch-off the converter. The OVP fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF. Although the converter has latched-off in response to an OVP fault, the LGATE gate-driver output will retain the ability to toggle the low-side MOSFET on and off, in response to
Where: - VO is the desired output voltage of the converter - The voltage to which the converter regulates the FB pin is the VREF - RTOP is the voltage-programming resistor that connects from the FB pin to the converter output. In addition to setting the output voltage, this resistor is part of the loop compensation network - RBOTTOM is the voltage-programming resistor that connects from the FB pin to the GND pin Choose RTOP value first, and calculate RBOTTOM according to Equation 9:
V REF * R TOP R BOTTOM = ---------------------------------V O - V REF (EQ. 9)
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Programming the PWM Switching Frequency
The ISL6228 does not use a clock signal to produce PWMs. The PWM switching frequency fSW is programmed by the resistor RFSET that is connected from the FSET pin to the GND pin. The approximate PWM switching frequency is written as Equation 10:
1 f SW = -------------------------K R FSET 1 R FSET = ----------------K * f SW (EQ. 10)
General Application Design Guide
This design guide is intended to provide a high-level explanation of the steps necessary to design a single-phase power converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced in the following section. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts.
Estimating the value of RFSET is written as Equation 11:
(EQ. 11)
Selecting the LC Output Filter
The duty cycle of an ideal buck converter is a function of the input and the output voltage. This relationship is written as Equation 13:
VO D = --------V IN (EQ. 13)
Where: - fSW is the PWM switching frequency - RFSET is the fSW programming resistor - K = 1.5 x 10-10 It is recommended that whenever the control loop compensation network is modified, fSW should be checked for the correct frequency and if necessary, adjust RFSET.
The output inductor peak-to-peak ripple current is written as Equation 14:
VO * ( 1 - D ) I PP = ----------------------------f SW * L (EQ. 14)
Compensation Design
Figure 6 shows the recommended Type-II compensation circuit. The FB pin is the inverting input of the error amplifier. The COMP signal, the output of the error amplifier, is inside the chip and unavailable to users. CINT is a 100pF capacitor integrated inside the IC, connecting across the FB pin and the COMP signal. RTOP, RFB, CFB and CINT form the Type-II compensator. The frequency domain transfer function is given by Equation 12:
1 + s * ( R TOP + R FB ) * C FB G COMP ( s ) = ------------------------------------------------------------------------------------------s * R TOP * C INT * ( 1 + s * R FB * C )
FB
A typical step-down DC/DC converter will have an IP-P of 20% to 40% of the maximum DC output load current. The value of IPP is selected based upon several criteria such as MOSFET switching loss, inductor core loss, and the resistive loss of the inductor winding. The DC copper loss of the inductor can be estimated by Equation 15:
P COPPER = I LOAD
2
*
DCR
(EQ. 15)
Where ILOAD is the converter output DC current. The copper loss can be significant so attention has to be given to the DCR selection. Another factor to consider when choosing the inductor is its saturation characteristics at elevated temperature. A saturated inductor could cause destruction of circuit components, as well as nuisance OCP faults. A DC/DC buck regulator must have output capacitance CO into which ripple current IP-P can flow. Current IPP develops a corresponding ripple voltage VP-P across CO, which is the sum of the voltage drop across the capacitor ESR and of the voltage change stemming from charge moved in and out of the capacitor. These two voltages are written as Equation 16:
V ESR = I P-P * E SR (EQ. 16)
(EQ. 12)
CINT = 100pF
RFB
CFB
RTOP EA COMP + REF ISL6228 RBOTTOM VO FB
FIGURE 6. COMPENSATION REFERENCE CIRCUIT
and Equation 17:
I P-P V C = ----------------------------8 * CO * f
SW
The LC output filter has a double pole at its resonant frequency that causes rapid phase change. The R3 modulator used in the ISL6228 makes the LC output filter resemble a first order system in which the closed loop stability can be achieved with the recommended Type-II compensation network. Intersil provides a PC-based tool (example page is shown later) that can be used to calculate compensation network component values and help simulate the loop frequency response.
(EQ. 17)
If the output of the converter has to support a load with high pulsating current, several capacitors will need to be paralleled to reduce the total ESR until the required VP-P is achieved. The inductance of the capacitor can cause a brief voltage dip if the load transient has an extremely high slew rate. Low inductance capacitors should be considered. A capacitor
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ISL6228
dissipates heat as a function of RMS current and frequency. Be sure that IP-P is shared by a sufficient quantity of paralleled capacitors so that they operate below the maximum rated RMS current at fSW. Take into account that the rated value of a capacitor can fade as much as 50% as the DC voltage across it increases.
MOSFET Selection and Considerations
Typically, a MOSFET cannot tolerate even brief excursions beyond their maximum drain to source voltage rating. The MOSFETs used in the power stage of the converter should have a maximum VDS rating that exceeds the sum of the upper voltage tolerance of the input power source and the voltage spike that occurs when the MOSFET switches off. There are several power MOSFETs readily available that are optimized for DC/DC converter applications. The preferred high-side MOSFET emphasizes low gate charge so that the device spends the least amount of time dissipating power in the linear region. Unlike the low-side MOSFET which has the drain-source voltage clamped by its body diode during turn off, the high-side MOSFET turns off with VIN - VOUT, plus the spike, across it. The preferred low-side MOSFET emphasizes low r DS(ON) when fully saturated to minimize conduction loss. For the low-side (LS) MOSFET, the power loss can be assumed to be conductive only and is written as Equation 20:
P CON_LS I LOAD r DS ( ON )_LS * ( 1 - D )
2
Selection of the Input Capacitor
The important parameters for the bulk input capacitance are the voltage rating and the RMS current rating. For reliable operation, select bulk capacitors with voltage and current ratings above the maximum input voltage and capable of supplying the RMS current required by the switching circuit. Their voltage rating should be at least 1.25 times greater than the maximum input voltage, while a voltage rating of 1.5 times is a preferred rating. Figure 7 is a graph of the input RMS ripple current, normalized relative to output load current, as a function of duty cycle that is adjusted for converter efficiency. The ripple current calculation is written as Equation 18:
2 2D 2 ( I MAX ( D - D ) ) + x I MAX ----- 12 I IN_RMS, NORMALIZED = ---------------------------------------------------------------------------------------------------I MAX
(EQ. 20)
(EQ. 18)
For the high-side (HS) MOSFET, the its conduction loss is written as Equation 21:
P CON_HS = I LOAD
2
*
Where: - IMAX is the maximum continuous ILOAD of the converter - x is a multiplier (0 to 1) corresponding to the inductor peak-to-peak ripple amplitude expressed as a percentage of IMAX (0% to 100%) - D is the duty cycle that is adjusted to take into account the efficiency of the converter which is written as:
VO D = ------------------------V IN EFF (EQ. 19)
r DS ( ON )_HS * D
(EQ. 21)
For the high-side MOSFET, the switching loss is written as Equation 22:
V IN * I VALLEY * t ON * f V IN * I PEAK * t OFF * f SW SW P SW_HS = ---------------------------------------------------------------- + -----------------------------------------------------------2 2 (EQ. 22)
Where: - IVALLEY is the difference of the DC component of the inductor current minus 1/2 of the inductor ripple current - IPEAK is the sum of the DC component of the inductor current plus 1/2 of the inductor ripple current - tON is the time required to drive the device into saturation - tOFF is the time required to drive the device into cut-off
In addition to the bulk capacitance, some low ESL ceramic capacitance is recommended to decouple between the drain of the high-side MOSFET and the source of the low-side MOSFET.
NORMALIZED INPUT RMS RIPPLE CURRENT 0.60 0.55 0.50 0.45 0.40 0.35 0.30 0.25 0.20 0.15 0.10 0.05 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 x=1 x = 0.75 x = 0.50 x = 0.25 x=0
Selecting The Bootstrap Capacitor
The selection of the bootstrap capacitor is written as Equation 23:
Qg C BOOT = ----------------------V BOOT (EQ. 23)
Where: - Qg is the total gate charge required to turn on the high-side MOSFET - VBOOT, is the maximum allowed voltage decay across the boot capacitor each time the high-side MOSFET is switched on
DUTY CYCLE
FIGURE 7. NORMALIZED RMS INPUT CURRENT
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As an example, suppose the high-side MOSFET has a total gate charge Qg, of 25nC at VGS = 5V, and a VBOOT of 200mV. The calculated bootstrap capacitance is 0.125F; for a comfortable margin, select a capacitor that is double the calculated capacitance. In this example, 0.22F will suffice. Use an X7R or X5R ceramic capacitor. EN (Pins 11 and 24), and PGOOD (Pins 7 and 28) These are logic signals that are referenced to the GND pin. Treat as a typical logic signal. OCSET (Pins 10 and 25) The current-sensing network consisting of ROCSET and CSEN needs to be connected to the inductor pads for accurate measurement. Connect ROCSET to the phasenode side pad of the inductor, and connect CSEN to the output side pad of the inductor. Connect the OCSET pin to the common node of node of ROCSET and CSEN. FB (Pins 8 and 27), and VO (Pins 9 and 26) The VO pin is used to sense the inductor current for OCP. Connect the VO pin to the output-side of CSEN through resistor RO. The input impedance of the FB pin is high, so place the voltage programming and loop compensation components close to the VO, FB, and GND pins keeping the high impedance trace short. FSET (Pins 1 and 6) This pin requires a quiet environment. The resistor RFSET and capacitor CFSET should be placed directly adjacent to this pin. Keep fast moving nodes away from this pin. LGATE (Pins 16 and 19) The signal going through this trace is both high dv/dt and high di/dt, with high peak charging and discharging current. Route this trace in parallel with the trace from the PGND pin. These two traces should be short, wide, and away from other traces. There should be no other weak signal traces in proximity with these traces on any layer. BOOT (Pins 14 and 21), UGATE (Pins 13 and 22), and PHASE (Pins 12 and 23) The signals going through these traces are both high dv/dt and high di/dt, with high peak charging and discharging current. Route the UGATE and PHASE pins in parallel with short and wide traces. There should be no other weak signal traces in proximity with these traces on any layer.
Layout Considerations
As a general rule, power should be on the bottom layer of the PCB and weak analog or logic signals are on the top layer of the PCB. The ground-plane layer should be adjacent to the top layer to provide shielding. The ground plane layer should have an island located under the IC, the compensation components, and the FSET components. The island should be connected to the rest of the ground plane layer at one point.
VIAS TO GROUND PLANE GND OUTPUT CAPACITORS SCHOTTKY DIODE LOW-SIDE MOSFETS INPUT CAPACITORS
VOUT INDUCTOR HIGH-SIDE MOSFETS PHASE NODE
VIN
FIGURE 8. TYPICAL POWER COMPONENT PLACEMENT
Signal Ground and Power Ground
The bottom of the ISL6228 TQFN package is the signal ground (GND) terminal for analog and logic signals of the IC. Connect the GND pad of the ISL6228 to the island of ground plane under the top layer using several vias, for a robust thermal and electrical conduction path. Connect the input capacitors, the output capacitors, and the source of the lower MOSFETs to the power ground plane. PGND (Pins 17 and 18) This is the return path for the pull-down of the LGATE low-side MOSFET gate driver. Ideally, PGND should be connected to the source of the low-side MOSFET with a low-resistance, low-inductance path. VIN (Pins 2 and 5) The VIN pin should be connected close to the drain of the high-side MOSFET, using a low resistance and low inductance path. VCC (Pins 3 and 4) For best performance, place the decoupling capacitor very close to the VCC and GND pins. PVCC (Pins 15 and 20) For best performance, place the decoupling capacitor very close to the PVCC and respective PGND pins, preferably on the same side of the PCB as the ISL6228 IC.
Copper Size for the Phase Node
The parasitic capacitance and parasitic inductance of the phase node should be kept very low to minimize ringing. It is best to limit the size of the PHASE node copper in strict accordance with the current and thermal management of the application. An MLCC should be connected directly across the drain of the upper MOSFET and the source of the lower MOSFET to suppress the turn-off voltage spike.
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ISL6228 Typical Performance
100 VIN = 8V 95 90 EFFICIENCY (%) EFFICIENCY (%) 85 80 75 70 65 60 0 1 2 3 4 IOUT (A) 5 6 7 8 VIN = 12V VIN = 19V 100 VIN = 8V 95 90 85 80 75 70 65 60 0 1 2 3 4 IOUT (A) 5 6 7 8 VIN = 12V VIN = 19V
FIGURE 9. CHANNEL 1 EFFICIENCY AT VO = 1.5V
FIGURE 10. CHANNEL 2 EFFICIENCY AT VO = 1.8V
VO1
VO1
FB1
FB1
PGOOD1
PGOOD1
PHASE1
PHASE1
FIGURE 11. START-UP, VIN = 12V, LOAD = 0.25, VO = 1.05V
FIGURE 12. SHUT-DOWN, VIN = 12V, IO = 10A, VO = 1.05V
VO1
VO1
PHASE1 PHASE1 VO2 PHASE2 PHASE2 VO2
FIGURE 13. CCM STEADY-STATE OPERATION,VIN = 12V, VO1 = 1.5V, IO1 = 3A, VO2 = 1.8A, IO2 = 4A
FIGURE 14. DCM STEADY-STATE OPERATION,VIN = 12V, VO1 = 1.5V, IO1 = 1A, VO2 = 1.8V, IO2 = 1A
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ISL6228 Typical Performance (Continued)
IO1 IO2
VO1
VO2
PHASE1
PHASE2
FIGURE 15. TRANSIENT RESPONSE, VIN = 12V, VO = 1.5V, IO = 0.1A/8.1A @ 2.55A/s
IO1
FIGURE 16. TRANSIENT RESPONSE, VIN = 12V, VO = 1.8V, IO = 0.1A/8.1A @ 2.55A/s
IO2
VO1
VO2
PHASE1
PHASE2
FIGURE 17. LOAD INSERTION RESPONSE, VIN = 12V, VO = 1.5V, IO = 0.1A/8.1A @ 2.55A/s
IO1
FIGURE 18. LOAD INSERTION RESPONSE, VIN = 12V, VO = 1.8V, IO = 0.1A/8.1A @ 2.55A/s
IO2
VO1
VO2
PHASE1
PHASE2
FIGURE 19. LOAD RELEASE RESPONSE, VIN = 12V, VO = 1.5V, IO = 0.1A/8.1A @ 2.55A/s
FIGURE 20. LOAD RELEASE RESPONSE, VIN = 12V, VO = 1.8V, IO = 0.1A/8.1A @ 2.55A/s
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation's quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 15
FN9095.2 May 7, 2008
ISL6228
Package Outline Drawing
L28.4x4A
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 0, 3/07
4X 2.4 4.00 A B 6 PIN 1 INDEX AREA 22 24X 0.40 28 1 6 PIN #1 INDEX AREA
21
4.00
2 .40 0 . 15
15 (4X) 0.15 14 8 0.10 M C A B 4 28X 0.20 28X 0.45 0.10
TOP VIEW
BOTTOM VIEW
SEE DETAIL "X" 0.10 C BASE PLANE ( 24X 0 . 4 ) ( 2. 40 ) SEATING PLANE 0.08 C
( 3. 75 TYP )
0 . 75
C
SIDE VIEW
( 28X 0 . 20 ) C 0 . 2 REF 5
( 28X 0 . 65)
0 . 00 MIN. 0 . 05 MAX.
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature.
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